Cancellation of spurious intermodulation products produced in nonlinear channels by frequency hopped signals and spurious signals

ABSTRACT

A method and apparatus for intermodulation product (IMP) cancellation. In one embodiment, the method comprises: acquiring copies of source signals that create IMPs in a passband of interest; creating copies of the IMPs for use as IMP cancellation signals by either multiplying the source signals together as a series of digital samples such that the multiplied signals create a near real and continuous time copy of the IMPs or creating a sum of the source signals in near real and continuous time and convolving the sum of the source signals with a mathematical model to effectively multiply the signals together to create a copy of the IMPs; adjusting one or both of phase and amplitude of the copies; and using the copies to cancel the IMPs inband of the passband of interest.

PRIORITY

The present patent application is a continuation of U.S. patentapplication Ser. No. 16/921,743, filed on Jul. 6, 2020 which is acontinuation of U.S. patent application Ser. No. 16/006,622, titled“Cancellation of Spurious Intermodulation Products Produced in NonlinearChannels by Frequency Hopped Signals and Spurious Signals,” filed onJun. 12, 2018 which is a continuation of U.S. patent application Ser.No. 15/165,993 titled “Cancellation of Spurious Intermodulation ProductsProduced in Nonlinear Channels by Frequency Hopped Signals and SpuriousSignals,” filed on May 26, 2016 and claims priority to and incorporatesby reference the corresponding provisional patent application Ser. No.62/167,072, titled, “Cancellation of Spurious Intermodulation ProductsProduced in Nonlinear Channels by Frequency Hopped Signals and SpuriousSignals,” filed on May 27, 2015, both of which are incorporated in theirentirety by reference.

FIELD OF THE INVENTION

The field of the invention relates to radio receivers and the mitigationof passive and active intermodulation products produced in nonlinearchannels by both continuous spurious and frequency hopped signals.

BACKGROUND OF THE INVENTION

When signals are transmitted wirelessly through a nonlinear channel,they will mix with each other, self-mix and or mix with noise componentsin the channel creating unwanted signal components that createinterference. Active intermodulation products are produced in the activecomponents, such as, for example, amplifiers and mixers, duringprocessing of wireless signal. Passive intermodulation products aregenerated after the amplifier by interaction with external materialssuch rusty components and external reflective surfaces.

FIG. 1 shows a typical example of an amplifier transfer curve. This typeof transfer function is also generally representative of any non-linearchannel. The 1:1 curve is known as the linear region of the amplifierand when the amplifier is driven beyond this range, the deviation fromlinear is experienced. The curved line is the compression curve of theamplifier. This can be modelled as a power series expansion such asaX+bX²+cX³+dX⁴ . . . and power series expansions like this predict thehigher order intermodulation products. All signals convolved with thistransfer function are effectively multiplied together and this createsIntermodulation Products. Intermodulation Products (IMP) are of theorder 2^(nd), 3^(rd), 4^(th) etc. In most cases, the even order IMPswill not fall inband, but if they do, they can be handled as describedherein for odd order IMPs.

INTERFERENCE ENVIRONMENT

When high power out of band signals (source signals) are incident on theRF front end of a wideband receiver such as a cell phone, or cell phonebase station, intermodulation products (IMPs) can be generated in theanalog components of the receiver. These intermodulation products (IMPs)can have significant power. This is the case for cellular handsets whichmust receive the entire telephone band because the receiver does nothave a prior knowledge of the channel to which it will be assigned. Thechannel selection is done after frequency down conversion becauseadjustable channel selection at RF is not practical and the insertionlosses would be unacceptable.

Passive IPMs are created in the microwave components and structuresafter the transmitter filter and are thus not removed by the systemfilters and can land inband of the receive signals.

In cellular base stations, selective filters provide for rejection ofadjacent channels that are not of interest, but passive IMPs can begenerated by signals from the individual service provider and or otherservice providers because the source signals are out of band of thepassband of interest, but the passive IMPs may not be and when they fallinband of the signal of interest, significant interference can berealized. The wideband LTE, Gen 4 and Gen 5 systems will make thepassive IMP problem even more severe.

SUMMARY OF THE INVENTION

A method and apparatus for intermodulation product (IMP) cancellation.In one embodiment, the method comprises: acquiring copies of sourcesignals that create IMPs in a passband of interest; creating copies ofthe IMPs for use as IMP cancellation signals by either multiplying thesource signals together as a series of digital samples such that themultiplied signals create a near real and continuous time copy of theIMPs or creating a sum of the source signals in near real and continuoustime and convolving the sum of the source signals with a mathematicalmodel to effectively multiply the signals together to create a copy ofthe IMPs; adjusting one or both of phase and amplitude of the copies;and using the copies to cancel the IMPs inband of the passband ofinterest.

BRIEF DESCRIPTION OF THE DRAWINGS

The present invention will be understood more fully from the detaileddescription given below and from the accompanying drawings of variousembodiments of the invention, which, however, should not be taken tolimit the invention to the specific embodiments, but are for explanationand understanding only.

FIG. 1 is a conceptual drawing of the nonlinear transfer function of anamplifier or a nonlinear channel capable of generating active and orpassive IMPS.

FIG. 2 is a system block diagram of an instantiation of an IMPcancellation process with frequency hopped source signals.

FIG. 3 is an example timeline for IMP cancellation with frequency hoppedsignals.

FIG. 4 illustrates 802.11b/g interference cancellation with frequencyhopped signal, with non-cooperative jamming signals.

DETAILED DESCRIPTION

Techniques described below enable the cancellation of IMPs in thereceiver, either passive or active IMPs, in continuous and near realtime. The techniques handle IMPs generated by continuous signals,spurious and those generated by frequency hopped signals. Spurious IMPscan be managed by the time delay implementation described herein as wellas the frequency hopped signals.

In one embodiment, the source signals that create the IMPs are capturedin the receiver and are then isolated with digital or analog filters,and these signals are then multiplied together to create a copy of theIMP to provide an IMP cancellation signal.

In one embodiment, the source signals (those that create the IMPs) aredigitally multiplied together sample by sample to create the IMPcancellation signal. As an example, if the sources signals are S1 andS2, then the cancellation signal is created by S1*S1*S2, S1*S1*S1, orS1*S2*S2 which creates IMP cancellation signals at 2F1-F2. 2F1-F1 and2F2-F1, where F1 is the center frequency of S1 and F2 is the centerfrequency of S2.

As one skilled in the arts will recognize, the convolution of the sum ofthe signals that create the IMP with the nonlinear model of an amplifierwill effectively multiply the signals together to create the IMPcancellation signals. This will create a multitude of IMPs and therequired IMP cancellation signals are the filtered off from the full setof IMPs to create the IMP signals inband of the signal of interest (SOI)for the cancellation process.

In another embodiment, a mathematical model of the amplifier ornonlinear channel shown in FIG. 1 (such as aX+bX²+cX³+dX⁴ . . . ) isused and the sum of signals S1 and S2 is convolved with the transferfunction of the amplifier/nonlinear channel to effectively multiply theS1 and S2 signals together. As one skilled in the arts will recognize,there are several mathematical models of the amplifier/nonlinear channelwhich will exhibit different compression curves that can be used withthe techniques described herein.

In one embodiment, the source signals are continuous, over at least ashort period of time at least, and the cancellation IMPs are generatedby multiplying the signals together or convolving the sum of the signalswith the transfer function of an amplifier or a composite of thenonlinear channel. The IMP cancellation signal(s) are then subtractedfrom the receive channel to cancel the IMPs generated by either theactive or passive IMP generation mechanisms.

In one embodiment, the output of the IMP cancellation process is crosscorrelated with the IMP cancellation signal(s) and the phase andamplitude of the IMP cancellation signal is adjusted to reduce, andpotentially minimize, the cross correlation, thereby optimizing the IMPcancellation process.

In one embodiment, the source signals are frequency hopped and the IMPcancellation process uses a short memory buffer to provide the IMPcancellation process to find the IMPs and set up the cancellationprocess.

In one embodiment, the full passband of frequency hopped signals issampled and treated on one signal, where the frequency hopped band isone signal and it is multiplied by a continuous signal or anotherfrequency hopped passband signal to create the IMP cancellation signal.This provides a “plethora” of time delays as a FIR filter withadjustable taps will do. In this embodiment, the sum of the signals isconvolved with the amplifier/nonlinear channel transfer function toeffectively multiply the source signals together to create the IMPcancellation signal(s).

In one embodiment, the IMPS are created in the channel as passive IMPsand not in the amplifier. In this case, the source signals aremultiplied together sample by sample or the sum of the signals isconvolved with a type of nonlinear transfer function such asaX+bX²+cX³+dX⁴ . . . to effectively multiply the source signals togetherto create the IMP cancellation signals. As one skilled in the arts willrecognize, there are a family of power series expansions that candescribe a compression curve as shown in FIG. 1 and this does notdetract from the general nature or application of this invention.

In one embodiment, the phase and amplitude of the IMP cancellationsignal is controlled with an adaptive FIR filter or an adaptive IIRfilter to accommodate not only linear phase and amplitude adjustment,but also to group delay to improve, and potentially optimize the IMPcancellation process.

In one embodiment, the phase and amplitude of the IMP cancellationsignals are adjusted individually as separate functions to reduce, andpotentially minimize, the cross correlation of IMPs process output andthe IMP cancellation signal to reduce, and potentially minimize thecross correlation and thus improve the IMP cancellation process.

In one embodiment, the IMP cancellation signal is filtered by a copy ofthe matched filter for the signal of interest (SOI) from which the IMPsignal is being removed. This provides for a filter matching of the IMPsin the SOI and the IMP cancellation signal.

In one embodiment, multiple IMPs are cancelled simultaneously as eithera sum of IMP cancellation signals or as a parallel cancellationprocesses.

In one embodiment, the source signals consist of several source signalsand when the sum of these signals is convolved with the amplifiertransfer function, the sum of the IMPs that fall in the signal ofinterest are filtered in an SOI matched filter and multiple IMPs arecancelled simultaneously.

When a source signal that creates an IMP goes way, the IMP cancellationsignal also goes away because one of the signals is effectively zero andwhen multiplying by zero, zero results. The process does contain arevisit mechanism to determine when an IMPs goes away so thecancellation process can get terminated so as not to add noise to thesignal path.

In the following description, numerous details are set forth to providea more thorough explanation of the present invention. It will beapparent, however, to one skilled in the art, that the present inventionmay be practiced without these specific details. In other instances,well-known structures and devices are shown in block diagram form,rather than in detail, in order to avoid obscuring the presentinvention.

Terms and Definitions

Channel: This is the electrical and radio frequency propagation path ofthe signals and includes the hardware components, atmosphere and freespace.

Active nonlinear device: physical device that contains some sort ofamplification and/or power translation, and in one embodiment amplifiersand mixers.

Passive Nonlinear Device: a physical device that creates nonlinearitieswithout amplification. These include, but are not limited to waveguides,feeds and other signal transmission channels wherein nonlinear effectsare experienced due to hardware element mismatch, oxidation and otherphysical degradations that cause nonlinear performance.

IMP: Intermodulation products: These are new signals created by thenonlinear effects in channels where signals self-mix and/or mix withother signals to create unwanted signals in the channels. In oneembodiment, these are modeled as a power series expansion such asaX+bX²+cX³+dX⁴ . . . and other power series expansions. These componentsare the result of signals being multiplied by themselves and othersignals. There may be and often are more than one IMP signal in thechannel,

IMB: Intermodulation distortion. This distortion seen in a channel whenactive signals are present and is manifested by intermodulation products(IMP)

Source Signals: These are the signals from which IMPs are created in thenonlinear channels, both passive and active.

Signals of Interest (SOI): These are the desired signals in the channeland they are often interfered with by the IMP signals. There may be oneor more SOIs in a channel.

Intercept points: As shown in FIG. 1, the intercept point is thelocation on the Input-Output curve of a nonlinear device where the 1:2,1:3, 1:4 lines intercept the 1:1 curve (line), In FIG. 1, the 3^(rd)order intercept point is shown, IP3.

Intermediate frequency (IF): Communications signals are often processedwithin the transmitter and receiver hardware at frequencies that differfrom the transmit frequencies. This is done for system performancereasons and the signals are converted to the Radio Frequency (RF) priorto transmission or from RF after reception.

Radio Frequencies (RFs): The radio frequencies are the frequencies ofthe signals that are used to transmit the signal thru the atmosphere orfree space or other channels such as waveguide.

I and Q channels: In analog and digital signal processing, passbandsignals are often separated into orthogonal channels referred to as Iand Q.

As one skilled in the arts will recognize, the methods described in theinvention are applicable to both even and odd IMPs.

Spurious IMPs

Spurious IMPs can be generated as either active or passive IMPs SpuriousIMPs can result from spurious signals in the transmit and/or receivepaths of the communications system. Spurious IMPs may not be present,but in one embodiment, the processors scan the receive environment forspurious IMPs an if they exist, the search algorithms search for thespurious source signals and delay the IMP cancellation and the SOI pathto allow time for the IMP cancellation signals to be generated and theIMPs in the passband of interest to be cancelled.

As shown in FIG. 1, the 1:3 line predicts the amplitude of the 3^(rd)order IMP given the input power of the source signals, and the inputthird order intercept point IP3 can be obtained with the techniquesdescribed herein, although it is not required for the IMD mitigation.The higher order intercepts points for the higher order distortions canalso be obtained. Additional lines, such as, for example, 1:5 and 1:7and 1:9 lines could also be drawn, but are omitted here to avoidcongestion. The methods described herein for generation of 3^(rd) orderIMPs are readily extendable to the higher order IMPs as one skilled inthe arts will recognize. The dominate IMP is normally the 3^(rd), so the3^(rd) is used herein for discussion and description of the invention.As one skilled in the arts will recognize, the methods used for 3^(rd)order IMPS are readily extended to the 5^(th), 7^(th), 9^(th) orderIMPs.

Wideband Multi-Signal Source Signals

In one embodiment, wideband signals or multiple signals comprise thesource signals in the band. In one embodiment, the entire ensemblesignals set is cubed, sample by sample, to create a composite 3^(rd)order IMP signal set. This set of signals is then filtered with a SOIfilter to only pass those IMP components that fall inband of the signalof interest. This filtered IMP cancellation signal is then used tocancel the IMPs inband of the SOI. The output of the cancellationprocess is cross correlated with the IMP cancellation signals and thephase, amplitude and group delay of the composite IMP cancellationsignal is adjusted to reduce the cross correlation and improve the IMPcancellation process. As one skilled in the art will recognize, thisprocess is extendable to 5^(th), 7^(th) and 9^(th) and higher order IMPs

The signal of interest (SOI) and the source signals that create IMPs canbe frequency hopped such as is optional in GSM, GPRS, EDGE and otherstandards and the frequencies can be hopped on the order of 0.5 msec ina pseudo random fashion. The IMP cancellation system provides thecapability to detect the interfering signals and cancels theinterference in near real time (NRT). This is accomplished byintroducing a small delay such as 0.1 msec delay and continuouslycancelling the interfering signals as will be discussed below. As oneskilled in the art will recognize, different delays in the architecture,to accommodate different frequency hopping rates, do not detract fromthe general nature an application of this invention.

The specified interference environment for CDMA2000 requires a two tonetest with the interfering signals at −21 dBm and the SOI at −94 dBm.When the interfering signals are not present, the sensitivityrequirement is −119.6 dBm. CDMA2000 gives up 25 dB in the presence ofhigh power intermodulation products (IMP). GSM, GPRS, EDGE and WCDMA setthe requirements for the source signals at −49 dBm. All of these systemscan operate in the same environment. The −49 dBm requirement assumesthat the handset is at least 250 meters from an offending base station.The frequency hopped option in GSM, GPSR and EDGE will mitigate thehigher power interferes to some degree especially for voice services,which can tolerate a lot of outages and still deliver acceptable, butnot totally desirable, performance. The data services may be impacted toa much higher degree.

System Overview

The following system overview is not intended to be a limiting case ofthe invention, but to serve as an illustrative example of the invention.

The example system, described in detail below, samples the entirereceive band with at low resolution (4 to 8 bits) and then conducts afast search to locate the energy which can produce an IMP within the SOIbandpass. Source signals that do cannot contribute to SOI inbandinterfering IMPs are not processed. The source signals that do havesufficient power and proper frequency spacing as required to produce aninband IMP are isolated and used to generate an estimate of the inbandIMP. The estimate of the IMP is generated by multiplying the sourcesignals together or are put through a nonlinear model that approximatesthe nonlinear elements that produce the IMPs. As one skilled in the artswill recognize, the nonlinear model can be an approximation of anamplifier compression curve or multiplying the signals together. Bothare nonlinear models. After appropriate phase and amplitude adjustment,the estimate of the IMP is used to cancel the IMP within the SOI. Thesource signals may and often do fall outside the passband of interestand are rejected in and of themselves with adjacent channel rejectionfilters, but this does not help with IMPs when they fall inband.

The process path for the SOI is parallel to the IMP search and path forthe SOI is processed at much higher resolution. The cancellation processdoes not require a high degree of resolution. If the cancellationprocess in accurate to 3 bits, the system will realize 18 dB ofinterference cancellation. As one skilled in the art will realize, ifthe SNR on the source signals is good and more effective bits arerealized in the sampling process, the IMPs cancellation process may beaccurate to many more bits and the suppression of the IMPs will begreater.

IMP Cancellation with Frequency Hopped Jamming Signals System BlockDiagram

The system block diagram for the cancellation of frequency hoppedjamming signals is shown in FIG. 2. Each block will be discussed indetail along with architecture options. Referring to FIG. 2, differentline types (e.g., solid, dotted, etc.) are used to show different typesof signals. Solid lines are used to show a single composite I-Q channelat some IF or RF carrier frequency. In one embodiment, the IF is adigital IF. When the processing is done on the individual I and Qchannels, the signal path is shown in dotted and the dashed lines arecontrol lines. The individual I and Q channels are not shown to avoidcongestion in the drawing. As one skilled in the arts will recognize,the RF bandwidths, data rates and signal path delays in this example arefor explanation only and do not limit the application of this inventionto different RF bandwidths, data rates and delay times. In this example,the delay times of 0.1 msec are just examples. In applications, thesedelays, blocks 2009, 2024, 2032 are determined by the particular signalconfigurations and signal environments.

For the example discussion below, cellular signals are used fordiscussion. This architecture and the techniques described herein alsoapply to any frequency hopped signals such as military anti jam (AJ)signals.

A key here is the sampling of the full receive band to include thesource signals that create the IMPs. The source signals are isolated andprocessed in a nonlinear model in the IMPs estimation model in block2018. This nonlinear model takes one of two basic forms, a nonlinearchannel model or a process in which the sources signals are multipliedtogether to create an estimate of the IMPs. The SOI processing path andthe IMP estimation paths are delayed as required to allow time for thegeneration of the IMP cancellation signals and the phase and amplitudeadjustments to improve the cancellation process.

Note that in the following discussion, the terms “block” and “unit” maybe used interchangeably. Such blocks and units include processing logicthat may comprise hardware (circuitry, dedicated logic, etc.), software(such as is run on a general purpose computer system or a dedicatedmachine), firmware, or a combination of the three.

Antenna: Block 2001

In one embodiment, the antenna receives the full passband of theallocated receive channel and the transmission path for the handsettransmission. In GSM like systems, the communications channel is notfull duplex and the transmission is not done simultaneously with thereception. In half duplex systems, an RF switch is used to alternatebetween transmit and receive channels. In CDMA and WCDMA systems, thehandset operates in a full duplex operation and a duplexer is used toprovide transmitter to receiver isolation. If the source signals forIMPs fall outside the receiver passband, the IMP cancellation processsamples the environment prior duplexer 2002 to get copies of the sourcesignals.

Duplexer: Block 2002

Duplexer 2002 is used in full duplex operation to provide attenuation ofthe transmitter feed thru to the receiver channel. In the PCS band, thepaired transmitter and receiver channels are separated by 80 MHz and thetransmitter signal is attenuated by around 50 to 55 dB by duplexer 2002.The transmitter power can be as high as 30 dBm and with 55 dB ofattenuation the transmitter feed thru seen by the receive channel can beas high as −25 dBm. When there is a close in blocker, cross modulationcan be experienced in the LNA. The specified blocker power level can beas high as −30 dBm. Due to the non-constant envelope on the transmitterfeed thru, the cross modulation (which is a function of the IIP3 in theLNA) of the required IIP3 in LNA 2003 must be at least +7.6 dBm if thereis no intermodulation suppression. The required sensitivity is −116 dBmfor the SOI.

In one embodiment, duplexer 2002 also provides suppression of the out ofband signals for the receive channel. In systems that do not operate ina full duplex mode, an RF SAW filter 2004 may be included prior to LNA2003 to suppress out of band signals that would normally be eliminatedby duplexer 2002. When SAW filter 2004 is used before LNA 2003, the IIP2in LNA 2002 can create interference when high power signals self mixedand create a “close to DC” interfering signals.

LNA: Block 2003

LNA 2003 provides the initial amplification of the receive signal andthe noise figure of this block is a key component of the system noisefigure. The IIP2 and IIP3 of LNA 2003 can induce significantinterference when high power blocker signals are present and create IMPsin band of the signal of interest.

RF SAW Filter: Block 2004

RF SAW filter 2004 is a fixed center frequency device with good out ofband attenuation, typically on the order of 50 to 55 dB. For the PCSband, filter 2004 is 60 MHz wide. In the case of the PCS band, RF SAWfilter 2004 passes the entire 60 MHz of the allocated receive band. Thereceiver receives the entire band because the channel to which thehandset will be assigned is not known prior to entry into the cell.After the amplification by LNA 2003 and the filtering by RF SAW filter2004, the specific channel selection is done as soon as possible eitherat IF or baseband. RF SAW filter 2004 further reduces the transmitterfeed thru and provides additional out of band signal rejection.

Mixer: Block 2005

Mixer 2008, after the LNA 2003 and RF SAW filter 2004, down converts thesignal to either and IF or baseband. The local oscillator (LO) used forthe frequency down conversion process is tunable such that the assignedchannel is converted to either a fixed IF or baseband where the channelselection filters are located. In one embodiment, Mixer 2005 is anonlinear device and the IIP2 and IIP3 of mixer 2005 will contribute tothe system IIP2 and IIP3. In one embodiment, Mixer 2005 also includes anAGC and buffer amplifiers which can also contribute to the system IIP2and IIP3. Since Mixer 2005 is after RF SAW filter 2004, the interferingsource signal outside of the allocated receiver band are greatlyattenuated. In one embodiment, the function is performed for both the Iand Q channels.

Baseband Analog Filters: Block 2006

In a super heterodyne receiver, an IF SAW filter is used to perform theassigned channel selection. In a direct conversion system, analogbaseband filters 2006 are used to provide some attenuation of theblocking signal and limitation of the noise bandwidth prior to the A/Dconverters 2007. The on chip analog filters 2006 will typically involveamplification as well as filtering and these are active filters. Activefilters will have non-linear characteristics and the IIP2 and IIP3 willcontribute to the system level IIP2 and IIP3. In one embodiment, thisfunction is performed for both the I and Q channels

Sigma Delta A/D Converter: Block 2007

In one embodiment, sigma delta A/D converter 2007 over samples thesignal at 1 bit per sample with typically a second or third order loopand the signal of interest is isolated by decimating (SOI) filters 2008.Decimating filters 2008 can be of different bandwidths, thus providing amulti-mode capability. The A/D conversion is done at baseband to providethe optimal SNR since the QSNR of the SOI is inversely proportional tothe highest frequency in the sampled signal. A higher the IF signalresults in a lower QSNR and thus a lower effective number of bits. Inone embodiment, the required number of bits is 13 to 14 bits and thus aSNR on the sampling process of 80 to 84 dB is used. In one embodiment,this function is performed for both the I and Q channels.

Signal of Interest (SOI) Decimating Filters: Block 2008

The over sampling of the sigma delta sampler is traded for SNR in thedecimating filters. The varying bandwidths of the decimating filters2008, in this block, provide the multi-mode capability. In oneembodiment, this function is performed for both the I and Q channels.

Programmable Delay Buffer: Block 2009

Programmable delay buffer 2009 is a delay that delays the SOI for ashort period of time so the frequency hopped jamming signals can beacquired and cancelled in cancelled block 2010. In one embodiment,Buffer 2009 is a first in and first out buffer and the amount of delayis adjustable depending on the nature of the frequency hopped orspurious IMPs. A one time difference between the SOI path, blocks 2001to 2010 and the interference cancellation path 2013 to 2032 is a staticdelay. Once this delay is known, the phase and amplitude of the IMPestimate can be adjusted in a very short period of time. If required,this can be determined with timing calibration at power up. In oneembodiment, the buffer 2009 provides a programmable delay and a steadyoutput of the delayed signal of interest to interference cancellationblock 2010 to provide a continuous cancellation of the interference witha system delay of about 100 microseconds. In one embodiment, thisfunction is performed for both the I and Q channels. As one skilled inarts will recognize, the magnitude of the delay can be varied to adjustto the frequency hopping features without impact to the general natureand application of the techniques described herein.

Interference Cancellation Block: Block 2010

In one embodiment, interference cancellation block 2010 comprisesdigital adders in which the estimate of the interfering signal(s) IMPsare subtracted from the SOI to cancel the inband interfering signals,IMPs. The cancellation block will typically consist of severalcancellation signals being subtracted simultaneously and the output fromthe block goes to the baseband processor and complex correlator 2029which can determine the effectiveness of the cancellation process andcan then provide fine tuning of the phase and amplitude of the estimateof the IMP. In one embodiment, this function is performed for both the Iand Q channels.

Interference Cancellation Calibration Block: Block 2011

Blocks 2011 thru 2030 provide the process for the development of theinterference cancellation signals. In these blocks, the estimate of theIMP is delayed in a parallel path to the calibration path blocks 2011,2012 and 2020 thru 2022. When the phase and amplitude adjustment for theIMP estimate is determined, the offsets are passed to the delayed pathto correct the phase and amplitude of the IMP estimated to be used tocancel the delayed copy of the SOI in blocks 2009 and 2010. Block 2011takes a copy of the SOI digital signal prior to the programmable delaybuffer 2009. Block 2011 provides digital cancellation of theinterference with an estimate of the interfering signal. Complexcorrelator block 2012 provides control signals for the adjustment of thephase and amplitude of the IMP estimate to optimize the interferencecancellation calibration process. In one embodiment, this function isperformed for both the I and Q channels.

Complex Correlator: Block 2012

In one embodiment, Complex correlator 2012 receives two inputs, one fromthe cancellation block 2011 after the cancellation process and one fromthe IMP computation process after the phase and amplitude adjustmentblock and the I-Q decomposition and digital down conversion in Block2023. Complex correlator 2012 first performs a correlation on the phaseby correlating a 90 degree phase shifted copy of estimate and the postinterference cancellation SOI from 2011. In one embodiment, the value ofthe correlation is minimized by adjusting the phase in blocks 2020 and2021. These are discussed in detail below. When the correlation of thephase has been minimized, correlator 2012 correlates the estimate of theIMP from block 2023, without the 90 degree phase shift, with the postcancellation signal from 2011. The amplitude is then adjusted in block2022 to help minimize the correlation process. In one embodiment, thisfunction is performed for both the I and Q channels.

In one embodiment, the phase and amplitude and group delay of the IMPcancellation signal are adjusted in a FIR filter or an IIR filter. Thisprovides a multitude of delays or a plethora of delays as required toadjust the phase and amplitude of the cancellation signal to optimizethe cancellation process.

Flash A/D Converter: Block 2013

The receive signal is sampled at the output of LNA 2003, or after the RFSAW filter, 2004. If the transmitter feed thru is to be acquired viathis path, then the signal is sampled at the output of LNA 2003. Analternate method is to get a sample of the transmit signal from thetransmitter and use this signal to compute the estimate of the crossmodulation interference. In one embodiment, if the transmit feed thru isrecovered directly from transmit channel, the output is taken the outputof RF SAW filter 2004 which provides the function of an anti-aliasingfilter which limits the sampled signal to 60 MHz passband. In this case,the flash A/D 2013 samples the signal on the order of 150 MHz. As oneskilled in the arts will recognize, the passbands of the signal sets andthe A/D converters 2013 will be implementation specific and do notdetract from the general capability and application of the techniquesdescribed herein.

In one embodiment, if the transmitter feed thru is recovered from thereceive chain, the passband is 140 MHz because it covers the entiretransmit and receive bands. In the case of CDMA2000 and other fullduplex systems, the signal of interest in the handset transmit band isthe transmit feed thru from that handset. All other signals will besignificantly attenuated by free space loss and duplexer 2002. As oneskilled in the arts will recognize, the passbands of the signal sets andA/D converters 2013 will be implementation specific and do not detractfrom the general capability and application of the techniques describedherein.

In one embodiment, flash A/D converter 2013 is a 4 bit A/D converter. Ifthe interference cancellation process is accurate to 3 or 4 bits, theinterference cancellation will be 18 to 24 dB at 6 dB per bit. As oneskilled in the arts will recognize, the A/D resolutions given here arefor illustration purposes and differences in the A/D resolution do notdetract from the general capability and application of the techniquesdescribed herein. If, for any reason, more resolution is required, theanalog value is down converted to some convenient IF and the pass bandis then sampled by A/D converter 2013 with more bits of resolution. Thesource signals are recovered with decimating filters and this increasesthe QSNR and the effective number of bits.

If the transmitter feed thru is recovered from the receive path, theflash A/D converter 2013 samples the passband at 300 MHz. The output ofthe A/D converter 2013 goes to filter blocks 2014 and 2030. Blocks 2030thru 2032 process the transmitter feed thru. Filter 2014 is the input tothe 2^(nd) and 3^(rd) order IMP estimate generation. As one skilled inthe arts will recognize, the passbands of the signal sets and the A/Dconverters will be implementation specific and do not detract from thegeneral capability and application of the techniques described herein.

This is discussed in detail below. In one embodiment, this process indone on a single composite signal and not on the I and Q channels. Asone skilled in the arts will recognize, the processing of the signals aspassband signals or the processing of the signals at complex I-Qbaseband does not detract from general capability or application of thetechniques described herein.

Receive Band Decimating Filter: Block 2014

Filter 2014 takes the 300 MHz 4-bit digital sample and the signal isfiltered to pass just the 60 MHz receive pass band and the signal isdown sampled to 150 MHz 4-bit samples for further processing. The 150mega samples per second are sent to blocks 2015 and 2016. This isdiscussed in detail below. In the case where either the transmit feedthru is not an issue, such as in systems that are not full duplex, andsystems where in the transmitter signal is acquired by sampling theoutput of transmitter output filter 2014 is not required.

In one embodiment, this process in done on a single composite signal andnot on the I and Q channels. As one skilled in the arts will recognize,the A/D resolutions and the analog passbands given here are forillustration purposes and differences in the A/D resolution do notdetract from the general capability and application of the techniquesdescribed herein.

Programmable Buffer: Block 2015

The search process in block 2016 introduces a finite delay thatdetermines how fast the source signals can be identified. The delay inprogrammable buffer block 2015 provides a delay equal to the timerequired for the search process to find the source signals in thereceive band. By delaying a copy of the sampled signal until thefrequency locations of the source signals can be identified, thefrequency hopped source signals, and thus the frequency hopped IMPs, canbe processed in near real-time and the interfering signal can becancelled as they come up, and very little if any residual interferencewill be experienced by the SOI. If the initial 5 microseconds of theinterference can be accepted, then programmable buffer 2015 may not berequired. The output of the delay buffer is sent to decimating filtering2017 where the source signals are isolated and the digital sample rateis reduced to 10 MHz. In one embodiment, this is discussed in detailbelow.

In one embodiment, this process is done on a single composite signal andnot on the I and Q channels. As one skilled in the arts will recognize,the A/D resolutions and the analog passbands given here are forillustration purposes and differences in the A/D resolution do notdetract from the general capability and application of the techniquesdescribed herein. This also applies to the signal delay paths.

Source Signal Search Algorithm; Block 2016

In one embodiment, the receive pass band of 60 MHz is sampled at 4 bitsand 150 MS/s. A 128 Point FFT is computed to get a frequency resolutionof about 470 kHz. In one embodiment, the FFT requires 448 complexmultiplications and 896 complex additions. This results in 1792 4-bitmultiplies and 1972 4-bit additions The time to collect sufficientsamples at 470 MHz is approximately 2 to 3 microseconds. The speed ofthe 4 bit 128 point FFT computation can be made very fast by performingparallel computation. If the 128 point FFT requires 64 parallelbutterflies, this results in each path computing 3583/64 operations (4bit multiplications and additions), which results in 56 operations ineach path. If a clock rate of 50 MHz is used, the delay is approximately1 to 2 microseconds. The number of gates and the clock rate can betraded off in the final implementation. As one skilled in the arts willrecognize, the A/D resolutions and the analog passbands given here arefor illustration purposes and differences in the A/D resolution do notdetract from the general capability and application of the techniquesdescribed herein. This also applies to the resolution on the FFTs orother passband search algorithms.

When the FFT has been computed, the 128 components are evaluated andthose components with sufficient amplitude are processed to determinewhich ones have the required frequency spacing to produce an IMP in bandof the SOI. When a pair of source signals that will result in an inbandIMP are identified, the frequencies are passed in control signals to thedecimating filters in filter block 2017.

In one embodiment, the band is re-searched every 5 microseconds and thesearch algorithm selects the worse case interfering signals. The processcan process multiple interfering signals simultaneously. In the case ofprocessing multiple interfering signals simultaneously, the searchalgorithm process passes control signals to multiple copies of block2017. As one skilled in the arts will recognize, the time frame forresearching the passband is application dependent and variations in thetimeframes do not detract from the general capabilities and applicationsof the techniques described herein.

In the case of signals that have sufficient amplitude to create 2^(nd)order products with enough power to interfere with the SOI, the singlesignal is isolated with a single decimating filter. The only differencebetween the processing downstream of a 2^(nd) and a 3^(rd) order IMPprocessing path is that the 2^(nd) order IMP estimate is generated bysimply multiplying a signal by itself or squaring the samples. This canalso be done by passing the signal through a nonlinear model of thecompression curve for the nonlinearity.

In one embodiment, the decimating filters 2017 will also isolate anyblocking signal close enough to the SOI for the transmitter feed thru togenerate cross modulation products in the SOI. The output of decimatingfilter block 2017 is sent to the IMP estimation generation block 2018and the cross modulation processing block 2032. This process in done ona single composite signal and not on the I and Q channels.

Decimation Filters: Block 2017

The search algorithm identifies the FFT components that have sufficientenergy and the required frequency spacing to product inband SOI 2^(nd)and 3^(rd) order IMPs interfering signals. These control signals areused to determine the center frequency for the programmable decimatingfilters 2017. The bandwidth of decimating filters 2017 is on the orderof 450 kHz and these are filters with a few poles, typically on theorder of 3 or 4 poles. The generation of the IMP estimates that followrequire a high fidelity time domain copy of the source signals and sothe filtering in the frequency domain is benign. As one skilled in thearts will recognize, the process of passing one or more signals throughthe model of a nonlinearity can be extended to multiple signals togenerate higher order IMPs, or the multiple signals can be multiplied byeach other in different combinations to generate higher order IMPs. Themultiplication of multiple signals together is another instantiation ofa nonlinear model.

In one embodiment, the output to IMP estimate generation block 2018 iseither one or two signals at 10 MHz 4-bit samples with a bandwidth ofapproximately 450 to 480 MHz. There are two signals for 3^(rd) order IMPprocessing and one for 2^(nd) order IMP processing. As one skilled inthe arts will recognize, the process of passing one or more signalsthrough the model of a nonlinearity can be extended to multiple signalsto generate higher order IMPs, or the multiple signals can be multipliedby each other in different combinations to generate higher order IMPs.The multiplication of multiple signals together is another instantiationof a nonlinear model.

This process is performed by filters 2017 in done on a single compositesignal or on the individual I and Q channels and it is done at somedigital IF which will then be decomposed into I and Q and digitally downconverted to baseband or zero IF in a downstream process.

Decimating filters 2017 increase the QSNR of the source signal samplesby the ratio of the reduction in bandwidth. This increases the effectivenumber of bits and improves the accuracy and magnitude of thecancellation process.

IMP Estimate Generation: Block 2018

The source signals that have been isolated in decimating filter block2017 are input to IMP estimate generation 2018. In the case of the3^(rd) order IMP estimates, the two signals from decimating filter block2017 are multiplied together samples by sample with one value beingmultiplied twice as required to generate the appropriate 3^(rd) orderIMP. The 3^(rd) order components that fall outside of SOI will befiltered out in the SOI filter block 2019. As one skilled in the artswill recognize, the process of passing one or more signals through themodel of a nonlinearity can be extended to multiple signals to generatehigher order IMPs, or the multiple signals can be multiplied by eachother in different combinations to generate higher order IMPs. Themultiplication of multiple signals together is another instantiation ofa nonlinear model.

In one embodiment, the sum (composite of the source signals) isconvolved with the mathematical model of the amplifier or nonlinearchannel to effectively multiply the signals together and create theestimate of the IMPs cancellation signal(s).

In the case of the 2^(nd) order IMPs, if a signal has sufficient powerto create a significant 2^(nd) order IMP at baseband, the signal isisolated in the decimating filters in decimating filter block 2017 andthe signal is squared, sample by sample, to generate the 2^(nd) orderIMP. Any components that fall outside SOI are filtered out in SOI filter2019.

In one embodiment, the IMP estimate is generated from copies of the realtime source signals that generate the IMPs that are inband of the SOI.When one or more of the source signals go away, the cancellation signalsalso goes away because one of the signals that is multiplied to get theestimate goes to zero, and when one number is multiplied by zero thecancellation signal is zero.

This process in done on a single composite signal and or on theindividual I and Q channels or it is done at some digital IF which willthen be decomposed into I and Q and digitally down converted to basebandor zero IF in a downstream process.

SOI Filter: Block 1019

When the IMP estimate block multiplies the time samples of the sourcesignals together, there will be components generated that do not fallinto the band of the SOI. The SOI filter block performs filtering of IMPcomponents that do not fall in the band of the SOI.

This process in done on a single composite signal and not on theindividual I and Q channels and it is done at some digital IF which willthen be decomposed into I and Q and digitally down converted to basebandor zero IF in a downstream process.

Calibration Macro Phase Adjustment: Block 2020

The estimate of the IMP may not have the same delay as seen in the SOIpath. The macro delay is a first-in first-out buffer that provides fullsample delays. In one embodiment, the time delay is a phase shift. Whenmore actuate phase adjustment is required, this is accomplished in themicro linear phase adjustment in block 2021. Macro phase adjustmentblock 2020 receives control signal from correlator. Complex correlator2012 which computes the correlation between the SOI after thecancellation process and the cancellation IMP signal. In one embodiment,complex correlator 2012 uses a dither or similar type algorithm toadjust the phase delay to minimize the cross correlation. It may use aFIR filter with a minimization algorithm.

The value of the phase adjustment is sent continually to macro phaseadjustment block 2025 which provides the correction to the delayed IMPestimate from programmable buffer 2024 to align it in time with thedelayed copy of the signal of interest in programmable buffer 2009.

This process in done on a single composite signal or on the individual Iand Q channels or it is done at some digital IF which will then bedecomposed into I and Q and digitally down converted to baseband or zeroIF in a downstream process.

Calibration Micro Phase Adjustment: Block 2021

When the sampled signal is over sampled from a Nyquist point of view,full symbol delays can result in very large phase shifts on the order of45 to 90 degrees. In this case, the phase shift may not be of sufficientresolution to align the estimate of the IMP to cancel the interferencein the SOI. In this case, a weighted average of the consecutive symbolsis computed and the new value is mapped in time to the time slot foranother original symbol. This can be viewed as a small FIR filter withtwo or more taps. Micro phase adjustment block 2021 receives a controlsignal from correlator 2012 which computes the correlation between theSOI after the cancellation process and the cancellation IMP signal. Inone embodiment, correlator 2012 uses a dither or similar type algorithmto adjust the phase delay to minimize the cross correlation.

The value of the phase adjustment is sent continually to micro phaseadjustment block 2026, which provides the correction to the delayed IMPestimate from macro phase adjustment block 2025 to align it in time withthe delayed copy of the signal of interest in programmable buffer 2009.

This process in done on a single composite signal and or on theindividual I and Q channels or it is done at some digital IF which willthen be decomposed into I and Q and digitally down converted to basebandor zero IF in a downstream process.

Calibration Amplitude Adjustment: Block 2022

There is a possibility that the amplitude of the IMP estimate may nothave the correct amplitude to correctly and completely cancel theinterference. Amplitude adjustment block 2022 provides the ability toadjust amplitude of the IMP estimate under the control of the complexcorrelator 2012. Amplitude adjustment block 2022 receives control signalfrom the complex correlator 2012 which computes the correlation betweenthe SOI after the cancellation process and the cancellation IMP signal.In one embodiment, complex correlator 2012 uses a dither or similar typealgorithm to adjust the phase delay to minimize the cross correlation.In one embodiment, the phase and amplitude are adjusted with a FIRfilter providing a multitude of plethora of time delays

In one embodiment, the value of the phase adjustment is sent continuallyto Amplitude adjustment block 2027, which provides the correction to thedelayed IMP estimate from micro phase adjustment block 2026 to align itin time with the delayed copy of the signal of interest in programmablebuffer 2009.

This process in done on a single composite signal and/or on theindividual I and Q channels or it is done at some digital IF which willthen be decomposed into I and Q and digitally down converted to basebandor zero IF in a downstream process.

Calibration I and Q Decomposition & Digital Down Conversion: Block 2023

Up to this point in the process, the IMP estimate generation process hasbeen conducted on an I-Q composite signal at a digital IF frequency. Inthis block, a digital down conversion to digital baseband I and Q isdone and the phase alignment of the I-Q down conversion is done underthe control of complex correlator 2012. The I and Q components of theIMP estimate are sent to complex correlator 2012 and to interferencecancellation calibration block 2011. Complex correlator 2012 adjusts thephase of the I and Q complex down conversion to optimize theinterference cancellation process.

The control output of block 2023 is sent continuously to block 2028which uses the same phase offsets to perform the complex down conversionto the baseband I and Q.

Buffer for IMP Estimate: Block 2024

Programmable buffer 2024 provides up to 100 microseconds of delay forthe 4-bit IMP estimate. This programmable buffer is used to adjust thetime delay of the IMP cancellation signal with the SOI delay inprogrammable buffer 2009. This buffer delay is controlled by the complexcorrelator 2029. As one skilled in the arts will recognize, themagnitude of delay and number of bits a quantization is an example hereand does not detract from the general nature, capabilities andapplications of this invention. Each instantiation will be signal pathdependent.

This process in done on a single composite signal or on the individual Iand Q channels or it is done at some digital IF which will then bedecomposed into I and Q and digitally down converted to baseband or zeroIF in a downstream process.

Delayed IMP Estimate Macro Phase Delay: Block 2025

The macro delay or full sample delay is performed with a first-in andfirst-out buffer. The macro level delay calibration has been computedcalibration loop in the macro phase adjustment block 2020. The delayoffset is continuously received from macro phase adjustment block 2020and is used to delay the IMP estimate from programmable buffer 2024. Inone embodiment, this delay allows for the cancellation process toprocess the IMP estimate with a delay of 100 microseconds which is thetime required to calibrate the IMP cancellation process so that there isno loss in the cancellation process when frequency hopping interferingsignal come and go. As one skilled in the arts will recognize, themagnitude of delay and number of bits a quantization is an example hereand does not detract from the general nature, capabilities andapplications of this invention. Each instantiation will be signal pathdependent.

This process in done on a single composite signal or on the individual Iand Q channels or it is done at some digital IF which will then bedecomposed into I and Q and digitally down converted to baseband or zeroIF in a downstream process.

Delayed IMP Estimate Macro Phase Delay: Block 2026

The micro delay or sub sample delay is performed in an interpolationfashion as discussed in micro phase adjustment block 2021. The microlevel delay calibration has been computed in a calibration loop in microphase adjustment block 2021. The delay offset is continuously receivedfrom 2021 and is used to phase shift the IMP estimate from programmablebuffer 2024 after 2025. This phase adjustment allows for thecancellation process to process the IMP estimate with a delay of 100microseconds which is the time required to calibrate the IMPcancellation process so that there is no loss in the cancellationprocess when frequency hopping interfering signal come and go. As oneskilled in the arts will recognize, the magnitude of delay and number ofbits of quantization is an example here and does not detract from thegeneral nature, capabilities and applications of this invention. Eachinstantiation will be signal path dependent.

This process in done on a single composite signal or on the individual Iand Q channels or it is done at some digital IF which will then bedecomposed into I and Q and digitally down converted to baseband or zeroIF in a downstream process.

Delayed IMP Estimate Amplitude Adjustment: Block 2027

The amplitude adjustment is performed in 2027. The amplitude calibrationhas been computed in calibration loop in amplitude adjustment block2022. The amplitude correction is continuously received from 2022 and isused to adjust the amplitude of the IMP estimate from the programmablebuffer 2024 thru 2026. This amplitude adjustment allows for thecancellation process to process the IMP estimate with a delay of 100microseconds which is the time required to calibrate the IMPcancellation process so that there is no loss in the cancellationprocess when frequency hopping interfering signal come and go. As oneskilled in the arts will recognize, the magnitude of delay and number ofbits of quantization are examples here and does not detract from thegeneral nature, capabilities and applications of this invention. Eachinstantiation will be signal path dependent.

This process in done on a single composite signal or on the individual Iand Q channels or it is done at some digital IF which will then bedecomposed into I and Q and digitally down converted to baseband or zeroIF in a downstream process.

Delayed IMP Estimate I & Q Decomposition & Down Conversion: Block 2028

Block 2028 continuously receives the I-Q phase rotation control fromBlock 2023 to down convert the estimate of the IMP coherently with theSOI buffered in programmable buffer 2009. Block 2028 outputs the I and Qvalues of the IMP cancellation signals to block 2010 for cancellation ofthe interfering inband IMPs.

SOI and IMP Estimate Complex Correlator: Block 2029

Complex correlator 2029 provides the final calibration and qualitycontrol check on the interference cancellation process. Complexcorrelator 2029 operates in the same may described above with respect tocomplex correlator 2012. If there are any adjustments required tooptimize the process, they will be identified here and fine adjustmentcan be made via control to blocks 2024 thru 2028. FIG. 2 shows the toplevel block diagram, and the full implementation this function isperformed for both the I and Q channels.

Transmitter Feed Thru Decimating Filter: Block 2030

The frequency of the transmit signal is known and is readily extractedfrom the 4-bit per sample of the signal at the output of the LNA by theprogrammable decimating filter in 2030. The 300 MHz 4-bit signal isfiltered in filter 2030 and the sample rate is reduced to 10 MHz 4-bitsamples. The required copy of the transmitter signal can be acquired bysampling the transmit signal at the output of the high power amplifier(HPA) and sampling it with a low speed 10 MHz A/D, and if this is done,the sampled signal in 2013 is taken after the RF SAW filter and issampled at 150 MHz at 4 bits. As one skilled in the arts will recognize,the band passes of the signals, the number of bits of quantization anddelays are application dependent and this example in not intended tolimit the scope, applications or functions of this invention.

This process in done on a single composite signal or on the individual Iand Q channels or it is done at some digital IF which will then bedecomposed into I and Q and digitally down converted to baseband or zeroIF in a downstream process.

Cross Modulation Processing: Block 2031

If the application of the techniques described herein is not a fullduplex system, or if there is no close in blocker to create crossmodulation, the cross modulation processing stream can be used to cancelanother 2^(nd) or 3^(rd) order IMP. This applies to blocks 2031 thru2034.

The cross modulation interference results when a high power interferingor blocking signal is within the bandwidth of the transmit signal fromthe band edge of the receive signal. When a high energy signal isencounter in the search block 2016, the decimating filters 2017 willisolate this signal and send the signal to the Cross modulationprocessing block 2031. Cross modulation processing block 2031 performsthe same functions as the blocks 2018 and 2019. With the two signalsthat create the cross modulation interference, namely the transmitterfeed thru and the close in blocker, an estimate of the cross modulationinterference is created. This process in done on a single compositesignal or on the individual I and Q channels or it is done at somedigital IF which will then be decomposed into I and Q and digitally downconverted to baseband or zero IF in a downstream process.

Cross Modulation Programmable Delay Buffer: Block 2032

Programmable delay buffer 2032 performs the same function asprogrammable buffer 2024.

This process in done on a single composite signal or on the individual Iand Q channels or it is done at some digital IF which will then bedecomposed into I and Q and digitally down converted to baseband or zeroIF in a downstream process.

Cross Modulation Phase and Amplitude Adjustment: Block 2033

The phase and amplitude adjustment block 2033 performs the samefunctions at 2020 thru 2022 and 2025 thru 2027.

This process in done on a single composite signal and not on theindividual I and Q channels and it is done at some digital IF which willthen be decomposed into I and Q and digitally down converted to basebandor zero IF in a downstream process.

Cross Modulation I & Q Down Conversion to Baseband: Block 2034

Decomposition block 2034 performs the same functions as described indecomposition block 2028. At the output of 2034, the I and Q signals aresent to a complex correlator like complex correlator 2029 as well as tothe cancellation block 2010.

System Timing and Delay

FIG. 3 shows the system timing and delay provided by the architectureshown in FIG. 3. Point A in FIG. 3 corresponds to LNA 2003 in FIG. 2. Atthe output of LNA 2003, the SOI processing path and the IMP estimategeneration path have zero delay. The total delay in the IMP processingpath and the SOI path must be matched.

The IMP processing path has a 0.005 msec delay at Point B whichcorresponds to programmable buffer 2015 and which provides thecapability to set up the decimating filters 2017 so as to capture theentire interfering source signals. The delay corresponds to the searchwindow.

Point C in FIG. 3 corresponds to programmable buffer 2024 in FIG. 2,which is a programmable delay used to adjust the delay between the SOIpath and the IMP Cancellation Path.

Point D in FIG. 3 corresponds to programmable buffer 2009 in FIG. 2,which is a programmable delay used to adjust the delay between the IMPCancellation Path and the SOI path.

Point E in FIG. 3 corresponds to cancellation block 2010 in FIG. 2 wherethe time delayed SOI and the IMP cancellation signals are summedtogether to cancel the interference in the SOI. The 100 microseconddelay provides the capability to find the source signals that canproduce inband IMPs and cancel them for all source signals includingfrequency hopped signals. As one skilled in the arts will recognize, thedelay is dictated by the hopping rate for frequency hopped signals andis application dependent and does not detract from the general nature,capability and application of the techniques described herein.

Frequency Hopped Narrowband Signals Interfering with 802.11b/g, WiFi, inthe ISM Band

The mitigation of narrowband interfering signal on wideband spreadspectrum (802.11b) and OFDM (802.11g) involves quickly finding theinterfering signal, recovering a copy of the interfering signal andsubtracting the narrowband interfering signal from the SOI. When this isdone, the portion of the SOI that is within the bandpass of thenarrowband interfering signal will be lost. If the bandwidth of theinterfering signal is small compared to the SOI, the degradation in theSOI will be small. As an example, if a 22 MHz wide 802.11b signal hasinterference from a BT signal, the BT signal is 1 MHz wide and 1/22 ofthe 802.11b energy is lost. This will result in a SNR degradation of 0.2dB and this will be for a very short period of time. This same principleapplies for any signal that is narrowband compared to the SOI to includemicrowave ovens. The system tradeoff is the impact of the interferenceversus the SNR after interference cancellation which is equal to theratio of the bandwidths. The impact of WiFi on the BT signal is notconsidered to be a serious problem when they are not co-located. ThisSNR in not suffered when the WiFi (802.11b,g) and the BT are collocatedin a single computer as discussed below.

The ISM band is an unlicensed band which is shared by many applicationsincluding Bluetooth (BT) which is frequency hopped in a pseudo randomfashion. When the BT is collocated with the WiFi, the BT and WiFitransmission signals that produce the BT and WiFi receiver interferenceare readily available and are used to cancel the WiFi from the BT andthe BT from the WiFi. In the case of BT interference on WiFi fromsources that are not collocated, a different approach is used.

When mitigating the interference from non-collocated BT interference,the same delay technique described for the telephony above is used. Thetop level architecture diagram is shown in FIG. 4. In this architecture,all processing is done on the composite IQ signal at an IF frequencyeither digitally or in analog. After all interference mitigation hasbeen performed, the signal is converted to I-Q format with a complexdown conversion in the digital domain. This is done as part of thecancellation block 4008 which outputs the baseband digital I and Q.

While there can be a number of different interfering signals in the ISMband, the focus here will be on frequency hopped interfering signalsbecause continuous or near continuous interfering signals can be handledby the same architecture or a subset thereof.

The key frequency hopped interfering signal will be BT which is hoppedover 79 frequencies of the ISM band at 1600 times per second in a pseudorandom manner. Due to the pseudo random nature and short duration of thehops, the cancellation process will induce a 100 microsecond delay tofind the interfering signal and isolate it with a programmabledecimating filter as was done for the telephony. In this case, however,the source signal is the interfering signal and it can be used directlyto cancel the interference. The search algorithm is a little bitdifferent in that the components of the FFT are analyzed to find thosethat exceed the average by some to be determined value. This will bediscussed in detail below.

Antenna: Block 4001

The antenna receives the full ISM band. For WiFi and BT, a duplexer isnot required since the systems are not full duplex. For WiFi, there are22 MHz, (802.11b) and 20 MHz (802.11g) bands within the 84.5 MHz ISMband. There may be interfering signals such as microwave ovens, cordlessphones and BT signals also in the band which is unlicensed and anyonecan use it if they stay within specified power levels. The antenna willpass the signal of interest and all interfering signals in the ISM band.

Low Noise Amplifier LNA: 4002

LAN 4002 provides the initial amplification of the ISM band receivedfrom the antenna. IMPs are not typically the problem in WiFi because theISM band does not have high power transmitter base stations as seen incellular telephony systems. Users of the ISM however do see aconsiderable amount of co-channel interference, i.e. signals in the samefrequency channels. If the IIP2 and IIP3 do present a problem, they willbe handled in the same manner as described above. In some architectures,RF SAW filter 4003 may be placed prior to the LNA 4002 if there are highpower signals outside the ISM band that could produce 2^(nd) or 3^(rd)IMPs within the ISM band. There will be a higher insertion loss in thistype of architecture and the system noise figure will be higher.

RF SAW Filter: Block 4003

In one embodiment, RF SAW filter 4003 is an 84.5 MHz filter that filtersout of band signals and reduces the noise band width of the signal priorto input to mixer 4004.

Down Conversion Mixer: Block 4004

Mixer 4004 may contain an AGC and buffer amplifiers prior to mixer 4004,but they are not shown here for simplification. This architecture issingle conversion architecture as opposed to a direct conversionarchitecture as shown for the telephony applications. In thisarchitecture, the ISM band is down converted to some lower IF, possiblyhas low as 50 MHz. Any convenient IF can be used. The lower the IF, thehigher will be the SNR of the sampled signal. The Nyquist sampling ofthe WiFi signal is 44 MHz. If A/D converter 4005 samples the passband at200 MHz, decimating filters 4006 will result in a 200/44 gain in SNR ora factor of about 4.6, which is about 6.5 dB which will result in onemore effective bit over the sampling resolution. 802.11g requires about50 dB of instantaneous dynamic range, so the A/D converter for thisapplication will sample at 8 bits and the decimating filter will resultin a SNR of 55 dB or 9+ effective bits. If the application is only to beused for 802.11g, but only 802.11g, the resolution of the A/D can bereduced to about 4 bits. In one embodiment, the selection of the IF isthat it should be at or below 500 MHz to get a good SNR on the A/Dconversion.

Flash A/D Converter: Block 4005

As discussed above, the ISM band is down converted to an IntermediateFrequency (IF) so that the phase noise on the sampling clock does notdegrade the SNR to below required levels. An IF frequency below 500 MHzwill be adequate. Depending on the resolution required, Flash A/Dconverter 4005 will sample the passband at 4 to 8 bits at 200 MHz. Thesubsampling results in images of the ISM every 200 MHz and the desiredIF is digitally selected. The output of the A/D goes to three blockssimultaneously, namely 4006, 4011 and 4012.

802.11b/g Decimating Filters: Block 4006

The digital samples are filtered by a programmable decimating filter4006. This does the SOI channel selection for either the 802.11b (22MHz) or the 802.11g (20 MHz) and increases the bit resolution by 1 bit.In one embodiment, the sample rate is around 50 MHz at the output offilter 4006.

Blocks 4007 thru 4010

These block operate in the same manner as those described for blocks2009 thru 2012 only here the processing in done on the composite I-QSignal at a digital IF and not on the I and Q channels individually.

Search Algorithm

The search algorithm computes a 128 point FFT. It takes 2 to 3microseconds to collect the required snapshot of samples to compute theFFT which provides a frequency resolution of 84.5/128=0.66 MHz. Using 64parallel paths at 50 MHz, it takes 1 microsecond to compute the FFT asdescribed for Block 2016. In this search algorithm, the values of theFFT are compared to the average value, and narrowband jammers thatexceed a given threshold determine where the programmable decimationfilters will isolate the high power narrowband signals. Block 4012delays the samples from A/D converter and search Block 4011 sendscontrol signals to decimation filter 4012 to set the filters to isolatethe interfering signals from the BT or the other high power interferingsignals.

Programmable Buffer: Block 4012

Programmable buffer 4012 works like programmable buffer 2015 describedin FIG. 2. If the first 5 microseconds or so of the interference can betolerated, block 4012 can be eliminated.

Decimation Filter: Block 4013

Decimation filters 4013 work like the filters in decimation filter 2017.The difference here between the telephony and the WiFi applications isthat the signals isolated by decimation filters 4013 are thecancellation signals.

Calibration and Cancellation of Interference: Blocks: 4014 thru 4023

The calibration and cancellation of interference operate basically thesame way as the corresponding functions in the telephony. The differencehere is that all of the processing is done at a digital IF on acomposite I-Q signal and the decomposition to I and Q and final downconversion to baseband is done digitally in block 4008.

In a first example embodiment, a method comprises acquiring copies ofsource signals that create IMPs in a passband of interest; creatingcopies of the IMPs for use as IMP cancellation signals by eithermultiplying the source signals together as a series of digital samplessuch that the multiplied signals create a near real and continuous timecopy of the IMPs or creating a sum of the source signals in near realand continuous time and convolving the sum of the source signals with amathematical model to effectively multiply the signals together tocreate a copy of the IMPs; adjusting one or both of phase and amplitudeof the copies; and using the copies to cancel the IMPS inband of thepassband of interest.

In another example embodiment, the subject matter of the first exampleembodiment can optionally include adjusting group delay of one or moreIMP cancellation signals, and wherein adjusting the phase and amplitudeand group delay occurs with a finite impulse response filter (FIR) or aninfinite impulse response (IIR) filter.

In another example embodiment, the subject matter of the first exampleembodiment can optionally include that the IMPs are created by frequencyhopped source signals, and further comprising tracking the frequencyhopped generated IMPs using time buffering.

In another example embodiment, the subject matter of the first exampleembodiment can optionally include cross correlating an output of a IMPcancellation process with the IMP cancellation signal and then adjustingone or more of amplitude, phase and group delay of the IMP cancellationsignal to reduce cross correlation.

In another example embodiment, the subject matter of the first exampleembodiment can optionally include that the source signals are frequencyhopped signals, and further comprising processing the passband of thefrequency hopped signals and creating an IMP cancellation signal that iseither frequency hopped or continuous.

In another example embodiment, the subject matter of the first exampleembodiment can optionally include sampling a full passband to acquirethe source signals, the source signals creating IMPS in the passband ofinterest in a receiver.

In another example embodiment, the subject matter of the first exampleembodiment can optionally include that the copies of the source signalsare available from a co-located transmitter and copies of the sourcesignals are sent to a receiver and the IMP cancellation signals aregenerated in the receiver and passive and active IMPs are cancelled inthe receiver.

In another example embodiment, the subject matter of the first exampleembodiment can optionally include that source signals are acquired byeither or both sampling or from a co-located transmitter.

In another example embodiment, the subject matter of the first exampleembodiment can optionally include that the source signals are delayed anamount of time to allow for frequency hopped source signals to beidentified and isolated to create the IMP cancellation signals.

In another example embodiment, the subject matter of the first exampleembodiment can optionally include filtering at least one of the IMPcancellation signals, using a filter similar to a filter used on thesource signals, to pass only IMP components that will fall inband of thepassband.

In another example embodiment, the subject matter of the first exampleembodiment can optionally include that using the copies to cancel theIMPs inband of the passband of interest causes multiple IMPs to becancelled simultaneously by a parallel process with one IMP cancellationper process.

In another example embodiment, the subject matter of the first exampleembodiment can optionally include that using the copies to cancel theIMPs inband of the passband of interest causes multiple IMPs to becancelled simultaneously as a composite IMP signal of one or more IMPs,where the composite IMP cancellation signal set is generated byconvolving the sum of the source signals with a mathematical model of achannel for active and or passive IMPs.

In another example embodiment, the subject matter of the first exampleembodiment can optionally include that each of the IMP cancellationsignals becomes zero when one or more of the source signals is no longerpresent and only one source signal is left.

In another example embodiment, the subject matter of the first exampleembodiment can optionally include scanning source signals to determinewhen the IMPs are no longer present, and stopping the cancellationprocess so as not to inject noise as part of IMP cancellation.

In another example embodiment, the subject matter of the first exampleembodiment can optionally include that a passband of source signals iscontinuous, frequency hopped and spurious and is sampled and treated asa single source signal that is multiplied by other source signals tocreate a single or multiple IMPs cancellation signal set.

In another example embodiment, the subject matter of the first exampleembodiment can optionally include that a passband of source signals iscontinuous, frequency hopped and spurious and is sampled and treated asa single source signal that is convoluted with a mathematical model ofan amplifier or a nonlinear channel to effectively multiply the signalstogether to create an IMP cancellation signal set used to cancel IMPs inthe passband of interest. In another example embodiment, the subjectmatter of this example embodiment can optionally include filtering theIMP cancellation signals, with a filter similar to a filter used on thesource signals, to pass only IMP cancellation frequency components thatfall inband of the passband of interest.

In another example embodiment, the subject matter of the first exampleembodiment can optionally include that creating a copy of the crossmodulation IMP signal using a transmitter feed through to cancel crossmodulation IMPs in the passband. In another example embodiment, thesubject matter of this example embodiment can optionally includefiltering the cross modulation IMPs, with a filter similar to a filterused on the source signals, to pass only those IMP frequency componentsthat fall within the passband.

In a second example embodiment, an apparatus comprises an input toacquire copies of source signals that create IMPs in a passband ofinterest; a first processing path to create copies of the IMPs for useas IMP cancellation signals by either multiplying the source signalstogether as a series of digital samples such that the multiplied signalscreate a near real and continuous time copy of the IMPs or creating asum of the source signals in near real and continuous time andconvolving the sum of the source signals with a mathematical model toeffectively multiply the signals together to create a copy of the IMPs;an amplitude adjustment unit operable to adjust one or both of phase andamplitude of the copies; and a cancellation unit operable to use thecopies to cancel the IMPs inband of the passband of interest.

In another example embodiment, the subject matter of the second exampleembodiment can optionally include a finite impulse response filter (FIR)or an infinite impulse response (IIR) filter operable to adjust groupdelay of one or more IMP cancellation signals.

In another example embodiment, the subject matter of the second exampleembodiment can optionally include that the IMPs are created by frequencyhopped source signals, and further comprising a buffer to track thefrequency hopped generated IMPs using time buffering.

In another example embodiment, the subject matter of the second exampleembodiment can optionally include a cross correlator to perform crosscorrelating on an output of an IMP cancellation process with the IMPcancellation signal; and an adjustment unit to adjust one or more ofamplitude, phase and group delay of the IMP cancellation signal toreduce cross correlation.

In another example embodiment, the subject matter of the second exampleembodiment can optionally include that the source signals are frequencyhopped signals, and further wherein the processing path is operable toprocess the passband of the frequency hopped signals and create an IMPcancellation signal that is either frequency hopped or continuous.

In another example embodiment, the subject matter of the second exampleembodiment can optionally include a sampling unit to sample a fullpassband to acquire the source signals, where the source signals createIMPs in the passband of interest in a receiver.

In another example embodiment, the subject matter of the second exampleembodiment can optionally include that a co-located transmitter, whereinthe copies of the source signals are available from the co-locatedtransmitter and copies of the source signals are sent to a receiver andwherein IMP cancellation signals are generated in the receiver andpassive and active IMPs are cancelled in the receiver using acancellation unit.

In another example embodiment, the subject matter of the second exampleembodiment can optionally include that source signals are acquired byeither or both sampling or from a co-located transmitter.

In another example embodiment, the subject matter of the second exampleembodiment can optionally include a buffer to delay the source signalsby an amount of time to allow for frequency hopped source signals to beidentified and isolated to create the IMP cancellation signals.

In another example embodiment, the subject matter of the second exampleembodiment can optionally include a first filter to filter at least oneof the IMP cancellation signals to pass only IMP components that willfall inband of the passband, the first filter being similar to a secondfilter used on the source signals.

In another example embodiment, the subject matter of the second exampleembodiment can optionally include that the cancellation unit is operableto cancel multiple IMPs simultaneously via a parallel process with oneIMP cancellation per process.

In another example embodiment, the subject matter of the second exampleembodiment can optionally include that the cancellation unit is operableto cancel multiple IMPs simultaneously as a composite IMP signal of oneor more IMPs, where the composite IMP cancellation signal set isgenerated by convolving the sum of the source signals with amathematical model of a channel for active and or passive IMPs.

In another example embodiment, the subject matter of the second exampleembodiment can optionally include that each of the IMP cancellationsignals becomes zero when one or more of the source signals is no longerpresent and only one source signal is left.

In another example embodiment, the subject matter of the second exampleembodiment can optionally include a search unit to scan source signalsto determine when the IMPs are no longer present and to stop thecancellation process.

In another example embodiment, the subject matter of the second exampleembodiment can optionally include a sampling unit to sample a passbandof source signals that continuous, frequency hopped and spurious, thesignals processed as a single source signal that is multiplied by othersource signals to create a single or multiple IMPs cancellation signalset.

In another example embodiment, the subject matter of the second exampleembodiment can optionally include that a passband of source signals iscontinuous, frequency hopped and spurious and is sampled and treated asa single source signal that is convoluted with a mathematical model ofan amplifier or a nonlinear channel to effectively multiply the signalstogether to create an IMP cancellation signal set used to cancel IMPs inthe passband of interest. In another example embodiment, the subjectmatter of this example embodiment can optionally include a first filterto filter the IMP cancellation signals to pass only IMP cancellationfrequency components that fall inband of the passband of interest, thefirst filter being similar to a second filter used on the sourcesignals.

In another example embodiment, the subject matter of the second exampleembodiment can optionally include a cross modulation processing unit tocreate a copy of the cross modulation IMP signal using a transmitterfeed through to cancel cross modulation IMPs in the passband. In anotherexample embodiment, the subject matter of this example embodiment canoptionally include a first filter to filter the cross modulation IMPs topass only those IMP frequency components that fall within the passband,the first filter being similar to a second filter used on the sourcesignals.

Some portions of the detailed descriptions above are presented in termsof algorithms and symbolic representations of operations on data bitswithin a computer memory. These algorithmic descriptions andrepresentations are the means used by those skilled in the dataprocessing arts to most effectively convey the substance of their workto others skilled in the art. An algorithm is here, and generally,conceived to be a self-consistent sequence of steps leading to a desiredresult. The steps are those requiring physical manipulations of physicalquantities. Usually, though not necessarily, these quantities take theform of electrical or magnetic signals capable of being stored,transferred, combined, compared, and otherwise manipulated. It hasproven convenient at times, principally for reasons of common usage, torefer to these signals as bits, values, elements, symbols, characters,terms, numbers, or the like.

It should be borne in mind, however, that all of these and similar termsare to be associated with the appropriate physical quantities and aremerely convenient labels applied to these quantities. Unlessspecifically stated otherwise as apparent from the following discussion,it is appreciated that throughout the description, discussions utilizingterms such as “processing” or “computing” or “calculating” or“determining” or “displaying” or the like, refer to the action andprocesses of a computer system, or similar electronic computing device,that manipulates and transforms data represented as physical(electronic) quantities within the computer system's registers andmemories into other data similarly represented as physical quantitieswithin the computer system memories or registers or other suchinformation storage, transmission or display devices.

The present invention also relates to apparatus for performing theoperations herein. This apparatus may be specially constructed for therequired purposes, or it may comprise a general purpose computerselectively activated or reconfigured by a computer program stored inthe computer. Such a computer program may be stored in a computerreadable storage medium, such as, but is not limited to, any type ofdisk including floppy disks, optical disks, CD-ROMs, andmagnetic-optical disks, read-only memories (ROMs), random accessmemories (RAMs), EPROMs, EEPROMs, magnetic or optical cards, or any typeof media suitable for storing electronic instructions, and each coupledto a computer system bus.

The algorithms and displays presented herein are not inherently relatedto any particular computer or other apparatus. Various general purposesystems may be used with programs in accordance with the teachingsherein, or it may prove convenient to construct more specializedapparatus to perform the required method steps. The required structurefor a variety of these systems will appear from the description below.In addition, the present invention is not described with reference toany particular programming language. It will be appreciated that avariety of programming languages may be used to implement the teachingsof the invention as described herein.

A machine-readable medium includes any mechanism for storing ortransmitting information in a form readable by a machine (e.g., acomputer). For example, a machine-readable medium includes read onlymemory (“ROM”); random access memory (“RAM”); magnetic disk storagemedia; optical storage media; flash memory devices; etc.

Whereas many alterations and modifications of the present invention willno doubt become apparent to a person of ordinary skill in the art afterhaving read the foregoing description, it is to be understood that anyparticular embodiment shown and described by way of illustration is inno way intended to be considered limiting. Therefore, references todetails of various embodiments are not intended to limit the scope ofthe claims which in themselves recite only those features regarded asessential to the invention.

1-20. (canceled)
 21. An apparatus comprising: an input to acquire a fullbass-band of the transmitter; a processing path to convolve the fullbass-band of the transmitter with a nonlinear model of the transmitterpassband to create one or more composite IMP signals; and a cancellationunit operable to use the one or more composite IMP signals to cancelIMPs in the transmitter passband and out of band interference caused bya non-linear channel of the transmitter.
 22. The apparatus of claim 21wherein the composite IMP signal is a broadband IMP signal created bymultiplying the transmitter passband signal by itself two or more times.23. The apparatus of claim 21, wherein the processing path is performedat one location and the control and calibration signals for the one ormore composite IMP signals are passed to a receive signal of interestprocessing path.
 24. The apparatus of claim 21 wherein the processingpath is operable to perform frequency shifting one or more of thecomposite IMP signals to produce the simulated intermodulation products,adjusted to compensate for phase, frequency and amplitude differencesbetween the IMPs and the one or more one or more composite IMP signalsused for cancellation by the cancellation unit.
 25. An apparatuscomprising: an input to acquire copies of source signals; a firstprocessing path to create IMP cancellation signals in response to thesource signals, including compensating for differences between the IMPcancellation signals and IMPs in a band of a signal of interest byapplying frequency, amplitude and phase adjustments for variances infrequency, amplitude and phase; and a cancellation unit operable to usethe IMP cancellation signals to cancel IMPs in the signal of interest.26. The apparatus of claim 25 wherein the first processing path isoperable to determine the frequency, amplitude and phase adjustments bycross correlating one or more of the IMP cancellation signals with thesignal of interest and adjusts the frequency, amplitude and phase toreduce the cross correlation between an IMP in the band of the signal ofinterest and the IMP cancellation signals.
 27. A method comprising:creating two copies of a sampled passband using a digital copy of apassband containing one or more signals of interest and intermodulationproducts and the source signals that can produce active intermodulationproducts, a first copy of the two copies of the sampled passband beingused to pass a signal of interest and a second copy of the two copies ofthe sampled passband being used to reject the signal of interest andpreserve the source signals that can create active intermodulationproducts, the sample of the sampled passband having the source signalspotentially containing both continuous and spurious and frequency hoppedsignals; processing the passband that contains the source signalsthrough a mathematical model of nonlinearity to produce estimates ofactive intermodulation products in the source signals, the activeintermodulation products containing both continuous and intermittentsignals; filtering to obtain intermodulation products that fall in-bandof the signal of interest; cancelling the in-band intermodulationproducts using in-band estimates of the intermodulation products andintermodulation products cancellation signals (ICS); adjusting the phaseand amplitude of the estimate of the intermodulation products to alignthe in-band intermodulation product and ICS; and performing a crosscorrelation function in which the ICS is cross correlated with thesignal of interest after cancellation of the in-band intermodulationproducts to determine residual intermodulation products, whereinperforming the cross correlation comprises continuously adjusting thephase and amplitude of the ICS to minimize cross correlation.
 28. Amethod comprising: creating two copies of a sampled passband using adigital copy of a passband containing one or more signals of interestand intermodulation products and the source signals that can producepassive intermodulation products, a first copy of the two copies of thesampled passband being used to pass the signal of interest and a secondcopy of the two copies of the sampled passband being used to reject thesignal of interest and preserve the source signals that can createpassive intermodulation products, the sample of the sampled passbandthat has the source signals potentially containing both continuous andspurious and frequency hopped signals; processing the passbandcontaining source signals that potentially produce passiveintermodulation products (PIM) through a mathematical model ofnonlinearity to produce estimates of the active intermodulationproducts; filtering the estimates of the passive intermodulationproducts to pass intermodulation products that fall in-band of thesignal of interest; canceling the in-band passive intermodulationproducts (PIM) using in-band estimates of the passive intermodulationproducts (PIM) and passive intermodulation products cancellation signals(PIMC); adjusting phase and amplitude of the estimate of theintermodulation products to align the in-band intermodulation productand passive intermodulation cancellation signals (PIMC); and performinga cross correlation function including cross correlating the PIMCcorrelated with the signal of interest after cancellation of the in-bandpassive intermodulation products (PIM) to determine residualintermodulation products, wherein performing the cross correlationfunction comprises continuously adjusting the phase and amplitude of thePIMC to minimize the cross correlation.
 29. A method comprising:processing a source signal that covers a full passband of a signal ofinterest, wherein the source signal is the signal of interest or has asame passband as the signal of interest; multiplying the signal ofinterest or all combined signals in the passband to create a 3^(rd)order intermodulation signal centered at a center band of the signal ofinterest with a passband three times the bandwidth of the signal ofinterest, which constitutes out of band interference; multiplying thesignal of interest passband by itself three times to createintermodulation cancellation signal (ICS) for cancellation of the 3^(rd)order intermodulation signals centered in the passband of the signal ofinterest including the out of band interference; and convolving thesignal of interest with a mathematical model of the nonlinearity tocreate an ICS signal
 30. A method comprising: creating an estimate ofall the 3^(rd) order intermodulation products (IMPs) by cubing a fullpassband of a receiver to create all of the potential 3^(rd) order IMPSincluding the IMP caused by the signal of interest, wherein fullpassband includes all source signals that can create IMPs in thepassband of a signal of interest; multiplying the passband by itself 5,7 9 or 11 times to create higher order IMP cancellation signals (ICSs);filtering an estimate of all of the higher order IMPs to pass only thoseIMPs components that fall within a passband of the signal of interestand create creating an IMP cancellation signal (ICS); cancel the IMPsin-band of the signal of interest using the filtered estimate; andadjusting the ICS in phase and amplitude to improve the cancellationprocess.
 31. A method comprising: create an estimate of all the IMPs byconvolving a full passband of a receiver with a mathematical model ofthe nonlinearity (a power series expansion) to create all of thepotential IMPs including the IMP caused by the signal of interest,wherein the full passband includes all source signals that can createintermodulation products (IMP) in the passband of a signal of interest;filtering the estimate of all of the IMPs to pass only those IMPscomponents that fall within the passband of the signal of interest andcreate an IMP cancellation signal (ICS); cancelling the IMPs in-band ofthe signal of interest using the filtered estimate (ICS) as part of acancellation process; and adjusting the ICS in phase and amplitude toimprove the cancellation process.